1. Field of the Invention
The present invention relates to a frequency-mixing device. More particularly, the present invention relates to a frequency-mixing device and a frequency-mixing method of a direct-conversion receiver (DCR).
2. Description of the Related Art
As known in the art, an intermediate frequency (IF) is a frequency to which a carrier frequency is shifted as an intermediate step in transmission or reception. A zero-IF system directly converts a carrier signal to and from the baseband signal without any IF stages.
The zero-IF receiver-employs a direct-conversion technique, in contrast to that of a superheterodyne receiver which employs a dual-conversion technique. The superheterodyne architecture has been widely employed due to its excellent channel selectivity characteristics.
Because the zero-IF system utilizes one mixer stage to convert the carrier signal directly to and from the baseband without the need for a surface acoustic Wave (SAW) filter, the zero-IF system can save cost, weight, design space, and the system can be implemented on one chip.
There have been various attempts to use the zero-IF technique in a mobile communication system, such as GSM (Global System for Mobile Communication). Mobile communication systems employing the zero-IF technique have become widespread.
In particular, a direct-conversion receiver (hereinafter, referred to as “DCR”) adopting the zero-IF technique has the advantages of a simple circuit structure, low cost manufacture and smaller size as compared to those of a superheterodyne receiver. However, in DCRs, the second-order intermodulation distortion (IMD2) is a fundamental problem occurring in a frequency mixer included in the DCR. The IMD2 is due to a non-linearity of the frequency mixer that employs non-linear active elements. When an input signal ei is applied to a non-linear system, an output signal is generated as represented by Expression 1 below.eo=α0+α1ei+α2ei2+α3ei3+  [Expression 1]
where α1, α2, α3 represent first-, second- and third-order harmonic coefficients, respectively.
The output signal eo can be represented as a sum of sine waves. Various frequency signals are mixed with one another and then new frequency signals are generated according to Expression 1 in the non-linear system, which is an important characteristic of the non-linear system.
When input signal ei including two frequency components f1 and f2 or an input signal ei having two-tone are/is applied to a general non-linear circuit, other frequency components such as 2*f1, 2*f2, f1−f2, f1+f2, 3*f1, 3*f2, 2*f1−f2, 2*f2−f1, 2*f1+f2, 2*f2+f1 and so on, as well as the input frequency components f1 and f2, are generated due to a non-linearity of the non-linear circuit.
Typically, the other frequency components generated due to the non-linearity are removed by a filter. However, when the input frequency components f1 and f2 are similar or identical to each other, and a target frequency signal among output frequency signals belongs to the baseband frequency signal, the frequency component f1−f2 close to the baseband frequency signal are hardly removed by the filter. These frequency component signals interfere with one another between channels having a small frequency difference, or distortion effects occur as signals within a particular frequency band interfere with one another.
The frequency component, resulting from a second-order component (or a second power term) such as the f1−f2 component and f1+f2 component, is the so-called IMD2 (second-order intermodulation distortion) component.
In a system such as the DCR, the f1−f2 component is included in a pass band filter for filtering a target frequency signal.
A relationship between a degree of the IMD2 and an amplified degree of an input frequency can represent a linearity of a circuit of the DCR system. The degree of the linearity of the circuit of the DCR system is represented by an IP2 (second-order intercept point).
A power of the initial IMD2 signal increases to a power level of an output IMD2 signal faster than a power of the input frequency signal increases to a power level of a target output frequency signal. Consequently, at first, the power level of the initial IMD2 signal is less than the power level of the output frequency signal; however, ultimately, the power level of the initial IMD2 signal becomes equal to the power level of the target output frequency signal. The power point at which the power level of the IMD2 signal is identical with the power level of the target output frequency signal is called the IP2 (second-order intercept point).
The larger the IP2 is, the better the linearity is, because a high power level of the input frequency signal is required in order that the power level of the initial IMD2 signal is equal to the power level of the target output frequency signal. A receiver second-order intercept point (IIP2), which represents an IP2 in view of an input, is a parameter for zero-IF applications. An output second-order intercept point (OIP2), which represents an IP2 in view of an output, is another parameter for zero-IF applications.
Because the DCR shifts the target frequency signal to the baseband, the IMD2 signal that is generated by the frequency mixer and which is located in the baseband can degrade the performance of the DCR. Therefore, the DCR requires a frequency-mixing device or a frequency mixer having a high IP2 value (or a low IMD2).
FIG. 1 is a circuit diagram illustrating a conventional Gilbert cell mixer. The Gilbert cell mixer is a kind of a balanced active mixer (typically, the balanced active mixer has a differential output characteristic). Referring to FIG. 1, the Gilbert cell mixer includes an emitter coupled transistor pair Q1 and Q2 for inputting a radio frequency signal pair RF+ and RF−, degeneration resistors RE1 and RE2, Gilbert cell core transistors Q3, Q4, Q5 and Q6, pull-up resistors R1 and R2, and differential output nodes N01 and N02.
When an identical second-order harmonic component is generated at each of the differential output nodes N01 and N02, the second order harmonic components of both differential output nodes N01 and N02 are counterbalanced with each other by a common-mode removal characteristic. As a result, the second-order harmonic components can be removed.
However, the second-order harmonic components are not completely removed, because the differential output nodes N01 and N02 generate the second-order harmonic components that have mismatches in phases and amplitude of the second-order harmonic components. The phase and amplitude mismatches are caused by: a mismatch between the emitter coupled transistor pair Q1 and Q2, a mismatch between the degeneration resistors RE1 and RE2, a duty ratio characteristic of a local oscillator LO, a mismatch between the pull-up resistors R1 and R2, and a mismatch between input radio frequency signals RF+ and RF−. Unfortunately, it is impossible to match the differential characteristic perfectly by removing all of above-mentioned mismatches.
FIG. 2 is a circuit diagram illustrating a conventional sub-harmonic mixer. Operations of the sub-harmonic mixer shown in FIG. 2 are disclosed in U.S. Pat. No. 6,587,678.
According to U.S. Pat. No. 6,587,678, the sub-harmonic mixer shown in FIG. 2 can perform a direct conversion using first through fourth quadrature signals. The first through fourth quadrature signals have a phase difference of about 90 degrees with respect to each other and have a local oscillator frequency corresponding to a half radio frequency (RF) signal. In FIG. 2, the reference symbols L00, L090, L0180 and L0270 represent signals corresponding to the first and the fourth quadrature signals, respectively, processed by a pre-processor according to a method disclosed in U.S. Pat. No. 6,587,678.
According to a method disclosed in U.S. Pat. No. 6,587,678, the direct conversion also can be performed using the local oscillator LO that generates an oscillating frequency lower than the frequency of the radio frequency (RF) signal. However, in U.S. Pat. No. 6,587,678, performance degradation due to the IMD2 was not solved.